Published on: Mar 3, 2016
Transcripts - Nabhighperformanceindoortvantennarpt
High‐Performance Indoor VHF‐UHF Antennas:
Technology Update Report
15 May 2010
(Revised 16 August, 2010)
M. W. Cross, P.E. (Principal Investigator)
Emanuel Merulla, M.S.E.E.
Richard Formato, Ph.D.
National Association of Broadcasters
Science and Technology Department
1771 N Street NW
Washington, DC 20036
Mr. Kelly Williams, Senior Director
100 Jackson Road
Devens, MA 01434
Section Title Page
1. Introduction and Summary of Findings……………………………………………..3
2. Specific Design Methods and Technologies Investigated…………………..7
2.1 Advanced Computational Methods…………………………………………………..7
2.2 Fragmented Antennas……………………………………………………………………..22
2.3 Non‐Foster Impedance Matching…………………………………………………….26
2.4 Active RF Noise Cancelling……………………………………………………………….35
2.5 Automatic Antenna Matching Systems……………………………………………37
2.6 Physically Reconfigurable Antenna Elements………………………………….58
2.7 Use of Metamaterials in Antenna Systems……………………………………..75
2.8 Electronic Band‐Gap and High Impedance Surfaces………………………..98
2.9 Fractal and Self‐Similar Antennas………………………………………………….104
2.10 Retrodirective Arrays…………………………………………………………………….112
3. Conclusions and Design Recommendations………………………………….128
1.0 Introduction and Summary of Findings
In 1995 MegaWave Corporation, under an NAB sponsored project, developed a
broadband VHF/UHF set‐top antenna using the continuously resistively loaded printed
thin‐film bow‐tie shown in Figure 1‐1. It featured a low VSWR (< 3:1) and a constant
dipole‐like azimuthal pattern across both the VHF and UHF television bands.
Figure 1‐1: MegaWave 54‐806 MHz Set Top TV Antenna, 1995
In the 15 years since then much technical progress has been made in the area of
broadband and low‐profile antenna design methods and actual designs. These
improvements have been published in: technical textbooks, peer‐reviewed articles,
patents, government research and development reports, and seminar proceedings. As
a developer of advanced antenna systems, primarily for the U.S. government,
MegaWave constantly reviews these sources and acquires the latest computer based
EM simulation tools in order preserve its competitive advantage. In this project, this
knowledge was used to identify ten candidate design methods and technologies that
have the potential to materially improve the performance of indoor VHF‐UHF TV
antennas. This report describes each candidate and its potential to improve indoor
”set‐top” reception of DTV signals between 54 and 698 MHz.
Of course, it must be kept in mind that, while advanced design methods and actual
physical designs exist, so do the laws of electromagnetics. Maxwell’s equations have
resulted both in practical as well as, what Dr. R. C. Hansen humorously calls,
“Pathological Antennas”. These pathological designs are described in his most recent
textbook , especially in the area of electrically‐small and broadband designs. It is
instructive to apply these fundamental limitations to the problem at hand, the set‐top
 Hansen, R.C., “Electrically Small, Superdirective, and Superconducting Antennas,” Wiley, 2006
Consider that a half‐wavelength in the low VHF TV band varies between 9.2 and 5.6
feet; between 34 and 27 inches in the high VHF band and between 12.6 and 8.5 inches
in the UHF (470‐698 MHz) band. A dipole antenna whose physical length is less than its
wavelength divided by pi (λ/π) is considered to be an electrically “small” antenna (ESA).
ESAs unfortunately are characterized by narrow bandwidths and low gains. Assuming 2
to 3 feet as a maximum acceptable length for an indoor or set‐top antenna, it definitely
falls into the ESA category in the low VHF band. But, in addition to size constraints and
the resulting difficulty in obtaining acceptable performance from a single antenna over
the 54 to 698 MHz spectrum, there are other concerns. Indoor and set‐top antennas
are fundamentally disadvantaged due to building penetration losses and by proximity
to sources of manmade radio noise. The former effect is more pronounced at UHF and
the latter at low VHF channels. Both can have a significant deleterious effect on
antenna performance. This brief discussion highlights the difficult problems inherent in
designing efficient, high performance antennas for the indoor/set‐op TV environment.
Fortunately, emerging technologies may effectively address these concerns.
This report is organized as follows. Sections 2.1 through 2.10 contain summaries of
each advanced method and hardware technology identified as a potential candidate for
high‐performance indoor VHF‐UHF DTV antennas. Each section includes a list of
references and, in many cases, photographs and performance data for multiple
implementations of the technology that is described. Section 3 includes conclusions
and a conceptual design for a practical indoor/set top VHF‐UHF antenna system.
The authors evaluated each technology and arrived at the conclusions and design
concept after sorting the nine hardware candidates into three categories as follows:
Mature technologies that do not require any CE‐909‐A channel designator or
signal quality information from the DTV receiver:
o Fragmented Antennas (Section 2.2)
o Non‐Foster Impedance Matching (Section 2.3)
Mature technologies that do require channel and quality data from the
o Active RF Noise Cancelling (Section 2.4)
o Automatic Antenna Matching Systems (Section 2.5)
o Physically Reconfigurable Antenna Elements (Section 2.6)
Emerging technologies that show promise, but are not sufficiently mature or
practical at this time:
o Metamaterials (Section 2.7)
o Electromagnetic Band Gap (EBG) Materials (Section 2.8)
o Fractal/Self Similar Antennas (Section 2.9)
o Retrodirective Arrays (Section 2.10)
A common thread connects each of these technology areas: advanced computational methods.
Whether a particular technology is mature and immediately applicable or emerging and highly
speculative, various schemes for antenna design optimization are universally applicable and
described in Section 2.1. These methodologies apply to all of the candidate technologies
discussed in Sections 2.2 through 2.10, and accordingly was placed at the beginning of Section
2. If even one of the optimization algorithms described had been available during the
development of MegaWave’s 1995 broadband set top antenna, it is likely that markedly better
gain performance would have resulted, especially in the low and high VHF bands. Another
attractive and potentially very significant capability offered by optimization algorithms is the
possibility of discovering entirely new antenna geometries, rather than simply optimizing a pre‐
Table 1‐1 subjectively ranks the nine identified candidate hardware technologies (2.2 ‐ 2.10). A
score of 10 represents perfection. By maturity we mean how close to off‐the‐shelf a particular
technology’s hardware is and how well it basic principle of operation has been vetted in the
literature. The term SWAP refers to size/weight and power.
Maturity Vetted Risk Design
2.1 Adv. Comp.
N/A N/A 9 9 Very Low N/A N/A Applies to all
2.2 Fragmented Passive No 7 7 Low Low 9 Planar
2.3 Non‐Foster Active No 6 7 Low Moderate 7 Limited
2.4 Active Noise
Active Yes 7 6 Low High 3 Requires
Active Yes 7 7 Moderate High 7 Requires
Active Yes 6 7 Moderate High 6 Control of
2.7 Metamaterials Passive No 3 4 High High 8 Emerging/
2.8 EBG Passive No 5 6 High High 5 Inherently
Passive No 6 4 Moderate Moderate 8 Controversial
2.10 Retrodirective Active Yes 4 5 Very High Very High 2 Narrow ‐
Table 1‐1: Candidate Technologies Considered and Their Ranking
As an example of how advanced computational methods could be combined with an
advanced hardware technique, that does not require a CE‐909‐A interface, is described at
the end of Section 3 and summarized here.
Using the genetic algorithm described in Section 2.1.5 a fragmented antenna was designed
and combined with a non‐Foster‐matching circuit to provide a planar 54‐698 MHz dipole
approximately 13 by 13 inches with significantly better gain, especially in the 54‐88 and
174‐216 MHz bands, than the 1995 MegaWave/NAB set top antenna. Figure 1 shows the
broadband fragmented planar element’s design obtained after approximately 24 hours of
computational time on a PC. Details of the specific method used are in Section 2.2 of this
report. It is well matched across the UHF DTV band, but requires some passive matching in
the high VHF band (which would also serve as the band combiner) and the more robust
matching capability of the active Non‐Foster‐Matching technique, described in Section 2.3,
for the low‐VHF band.
Figure 1. 13 x 13 Inch Planar Fragmented Non‐Foster Matched VHF‐UHF Antenna
An omni‐directional version could also be designed. It should be stressed that the above is
included here only to illustrate the notion of combining advanced computational broadband
antenna element designs with emerging electronic antenna matching capabilities and that
other antenna element geometries are also possible, depending on the starting conditions,
trade space dimensions and performance goals provided to the optimizer.
The authors want to make clear that 90 percent of the techniques and ideas contained in
this study are the work of others, as published in the open literature and referenced
2.0 Specific Design Methods and Hardware Technologies Investigated
2.1 Advanced Computational Methods
Optimization methodologies abound, and they are extensively used in every
aspect of engineering design, in particular antenna design. Optimization algorithms are
useful in two ways. They can be used to optimize the design parameters for a user‐
specified antenna geometry (for example, element spacing, length and diameter in a
Yagi‐Uda array). They also can generate designs that are impossible to achieve
otherwise. In both cases, optimization involves meeting specific performance objectives
(typically, VSWR, gain, bandwidth, and so on).
Optimization algorithms have become progressively more important as the
limitations of classic analytical techniques have become progressively more apparent.
While the equations underlying electromagnetic theory are well understood and
accurately describe all electromagnetic phenomena, in most practical cases they cannot
be solved analytically or, oftentimes, even numerically. Designing better antennas
requires improved methodologies, and state‐of‐the‐art optimization algorithms have
proven very effective. There is no question that these techniques are applicable to the
set‐top antenna design problem, and that they should receive considerable attention in
future design activities.
There are many different optimization methodologies that fall into two broad
categories: analytical methods and heuristic methods. Analytical methods are based on
precise mathematical formulations of the optimization problem. Even though they may
be fundamentally numerical in nature, they involve standard mathematical operations
such as computing derivatives or evaluating integrals. Heuristic methods may involve
equations, but the equations are not the result of an analysis. Instead, they are offered
without “proof” based on the fact that they “work.”
Many optimization heuristics are Nature inspired. The steps an algorithm
performs to optimize an antenna are based, for example, on how bacteria forage for
food. As disparate as these entities may seem, there is a connection, at least in the
sense that bacteria finding a good food source is similar to finding an antenna with a
good gain‐bandwidth product. Optimization algorithms of this type are usually referred
to as “metaheuristics,” a term intended to emphasize that the method is both empirical
and conceptual in nature. Thus, an effective bacteria foraging algorithm can be
implemented in many different ways because the bacteria foraging metaheuristic simply
suggests an analogy to Nature that is implemented in a computer algorithm working on
an antenna problem. The metaheuristic thus is an algorithmic framework instead of a
list of steps or instructions.
Several Nature inspired metaheuristics are described. A brief summary of each
algorithm is provided, and several example antenna problems solved by a variety of
algorithms are discussed. The algorithms include Ant Colony Optimization (ACO),
Particle Swarm Optimization (PSO), Genetic Algorithm (GA), Simulated Annealing (SA),
Central Force Optimization (CFO), Invasive Weed Optimization (IWO), Intelligent Water
Drop (IWD) algorithm, and Bacteria Foraging Optimization (BFO). There are many other
optimization algorithms [for example, Space Gravitation Optimization (SGO), Integrated
Radiation Optimization (IRO)], but they have not been applied to antennas or antenna
Each of these algorithms, except one, is inherently stochastic because its Nature
inspired algorithmic model relies on randomness in its functioning. The underlying
equations contain true random variables whose values are computed from a probability
distribution and consequently cannot be known in advance. As a result, every time a
stochastic optimizer run is made, its results are different than the previous run even
when exactly the same run setup parameters are used. The performance of stochastic
optimizers is necessarily characterized statistically (for example, average values,
standard deviations). This may be a limitation in the utility of optimization algorithms if
they are used in a set‐top antenna on a real time basis. For example, a self‐structuring
antenna (SSA) must reconfigure itself in real time in response, for example, to a
The one algorithm that is not inherently stochastic is Central Force Optimization
(CFO) whose Nature inspiring metaphor is gravitational kinematics, the branch of
physics that deals with the motion of masses moving under the influence of gravity. The
underlying equations are Newton’s equations of motion, which are completely
deterministic. CFO analogizes these equations in “CFO space” by flying “probes” that
are similar to small satellites to search a decision space “landscape” for the maximum
(optimal) values of a function (for example, antenna gain as a function of element length
and polar angle). CFO has been applied to antenna design and network synthesis, and
tested against many recognized benchmark functions used to evaluate optimization
algorithms. It therefore may be especially useful for the set‐top antenna problem.
This section describes developments in antenna design optimization over the
past fifteen years or so that have been driven largely by the availability of progressively
more powerful computers. A plethora of new optimization algorithms have been
introduced and tested and are now in widespread use. The new antenna designs often
are non‐intuitive, occasionally even counter‐intuitive, but all share the common feature
of not being accessible in any other way. State‐of‐the‐art optimization algorithms can
effectively solve intractable problems that have no analytical solutions or are too
complex to apply traditional analytical techniques. These approaches are useful right
now in designing set‐top television antennas, and they will continue to be useful
whatever form future set‐stop systems take. Some of the more important and
interesting optimization algorithms are described here.
Optimization Methodologies. The problem of locating the maximum values of a
function is generally referred to as “multidimensional search and optimization.” As
pointed out above, any problem involving three or more design parameters (“decision
variables”) is a multidimensional problem, and simple methods such as plotting the
function to be maximized cannot be used. Methods for solving these problems fall into
two broad categories: analytical methods and heuristic methods. Analytical methods,
which involve computing derivatives and gradients, are of limited use, especially in the
complex landscapes associated with antenna design. Stringent performance
requirements in terms of bandwidth, radiation pattern, and standing wave ratio (SWR)
make antenna optimization problems particularly difficult because the landscape is
usually extremely multimodal with narrow resonances and often high sensitivity to
slight parameter variations. Heuristic optimization methodologies, which are inherently
numerical in nature, are effective in dealing with these issues, and consequently they
are considered here while analytical approaches are not.
An entire class of heuristic optimization algorithms are “Nature inspired”, and
these appear to be the most effective. A Nature inspired algorithm is a computer search
and optimization program whose function mimics some natural process. These
programs are described as being “metaphorical” because they analogize some natural
process without precisely modeling it. For example, “Ant Colony Optimization” (ACO) is
an algorithm that simulates (to some degree) the behavior of ants seeking food. Thus,
ACO is inspired by the metaphor of ant foraging. All such algorithms evolve a solution to
the optimization problem over a series of steps or iterations, and almost all such
algorithms are stochastic population‐based methodologies. An initial population (of
ants, for example) randomly (stochastically) moves through the decision space
(landscape) step‐by‐step (iterating) in such a way that it converges on the largest food
supply (maximum function value). The ants’ progress is controlled by a set of equations
that mimic real ant behavior in Nature. There are many Nature inspired algorithms,
ACO being one of the earliest ones. The more important algorithms are discussed below
with examples of their application to antenna optimization.
2.1.3 Ant Colony Optimization
Figure 1 illustrates the basic idea behind Ant Colony Optimization (ACO) . The
irregular objects represent the ants’ nest (bottom) and a desirable source of food (top).
It has been observed that ants seeking food eventually traverse the shortest path
between the nest and food by marking that trail with a chemical pheromone that each
ant can sense (probably by smell). If the path is unobstructed [(a) in the figure], then
the ants simply walk a more‐or‐less straight line between home and the food supply.
But, if an obstruction is imposed [(b) and (c) in the figure], then more ants eventually
end up on the shorter trail between the food and the nest, which in turn results in a
greater pheromone concentration along that “optimal” trail. By depositing
progressively more pheromone on the shortest path, almost all of the ants eventually
end up on that path, and the “best” solution has been found. The red lines in the
bottom part of the diagram illustrate the path evolution with the eventual result that
the shortest path is identified.
The ACO algorithm mimics the ants’ behavior using equations that represent the
random motion of individual ants subject to their pheromone environment. Instead of
searching for food, the metaphorical ACO ants search the landscape of a decision space
for the maximum value of the function to be maximized. But the process they follow is
a simplified model of ant behavior as observed in Nature. And, just as real ants
eventually discover the best food source, ACO’s “ants” eventually converge on the
function’s global maximum value.
Figure 1. Ant Colony Optimization Metaheuristic (reproduced from ).
2.1.4 Particle Swarm Optimization
Particle Swarm Optimization (PSO)  is another stochastic population‐based
Nature inspired evolutionary algorithm. PSO analogies the swarming behavior of fish or
bees seeking food. Unlike ACO in which each “ant” creates a pheromone trail for other
ants to follow, PSO’s population of “agents” collectively communicate two pieces of
information: each individual agent’s “best” solution (greatest food concentration) and
the population’s overall (global) best solution. Equations that mimic bee and fish
swarming then control each agent’s subsequent motion in the decision space based on
the competing tendencies of moving toward the global best and randomly exploring the
vicinity of its best solution. As shown in Figure 2 for bees swarming around a flower
concentration, after many steps PSO agents converge on the global best solution
(highest flower concentration) because the local search fails to reveal any better
Figure 2. Particle Swarm Optimization metaheuristic (reproduced from ).
2.1.5 Genetic Algorithms
A Genetic Algorithm (GA)  analogizes the process of natural evolution or
“survival of the fittest.” When biological parents “mate,” they exchange DNA to create
a new individual (“child”) whose characteristics are drawn from both parents by
combining the parents’ DNA. A GA creates successive generations of children who then
serve as parents for the next generation whose children, in turn, will exhibit better
“fitness” than the previous generation. In the context of search and optimization, the
fitness is the value of the function to be maximized, so that the “best” fitness
corresponds to the function’s global maximum. As the GA progresses generation after
generation, the best discovered fitness improves and eventually converges on the
function’s global maximum.
Figure 3 shows a typical GSA flowchart for an antenna optimization algorithm. It
starts with a definition of the decision space (parameters to be optimized) and the
“fitness function” to be maximized (for example, antenna directivity, or some specified
combination of performance parameters such as gain, bandwidth, and so on). An initial
population of “individuals” is randomly created, and each individual is defined by a
chromosome that may be a binary sequence or a real number. Each chromosome
comprises a set of genes, and each gene is one of the design parameters. For example,
if the three design parameters were element length, inter‐element spacing, and
element diameter in a four element Yagi‐Uda array, then there is a total of eleven
design parameters, and each one is a gene. Thus, the optimization problem is defined
on an 11‐dimensional decision space, and the objective is to determine each of the
eleven parameters so as to maximize some specific fitness function, say, the array’s
gain‐bandwidth product. A separate computer program is used compute the fitness at
each step for each chromosome (the “evaluate fitness” box in Figure 3).
After the initial population’s fitnesses are evaluated, the “selection” process
chooses two parent chromosomes that will mate (“crossover”) to produce two children
chromosomes in the next generation. The selection and crossover processes take many
varied forms. For example, the selection of parents may be random, or “best mates
worst,” or best pairs pair wise through the population, and so on. The crossover
operation likewise can take many forms. For example, the parents’ chromosomes may
be split at the midpoint with first and second parts being swapped, or a random break
point might be used, or some other combinatorial approach taken. Finally, the children
thus created are subject to some level of mutation, a random perturbation of the
chromosome structure just as real chromosomes are mutated in Nature. The steps
described thus far are essentially common to all Gas, but the next step in the flowchart
(“elitist model”) is not. In this GA, the worst individual in the new generation is replaced
by the best individual from the previous generation, thus preserving the best solution
from generation to generation as the algorithm progresses. As a final step, the best
fitness is tested for convergence, and the process repeated until convergence is
Figure 3. GA flow chart (reproduced from ).
2.1.6 Simulated Annealing
Simulated Annealing (SA)  is a stochastic algorithm based on a metaphor
drawn from physics instead of biology, as ACO, PSO, and GA are. SA analogizes the
statistical mechanics of physical systems in thermal equilibrium with many degrees of
freedom. In particular, the physical processes involved in annealing a solid as it cools
forms the basis of the SA optimization algorithm, which has proven effective in
optimizing problems with large numbers of decision variables. Because of SA’s
complexity, the algorithm is not described in detail. Instead its performance against a
classic test problem is discussed.
The Traveling Salesman Problem (TSP) is a recognized example of combinatorial
optimization that SA was used to solve because it constitutes a good test of an
algorithm’s effectiveness. The salesman must visit N different cities once each and
return to his starting point. The problem is to determine the least costly route using a
“cost” or “objective” function that is specified beforehand. Minimizing the cost is the
same as maximizing its negative (note that minimization and maximization problems are
exactly the same except for multiplying the objective function by ‐1). The TSP is a
multidimensional search and optimization problem in the same vein as an antenna
optimization problem, so that an algorithm suitable for one very likely is applicable to
For the SA test, the TSP cost function is simply the total distance travelled by the
salesman (to be minimized). Two different distance metric can be used, the standard
Euclidean distance (“square root of the sum of the squares”), or the “Manhattan” metric
(sum of the separations along the two coordinate axes), the latter being used in this
case because it is simpler (less computationally intensive). Evolved solutions for TSP
appear in Figure 6 and show a clear tendency towards removing redundancy in the
travelled route, with the final solution (d) being close to optimal as discussed in .
Figure 4. Evolution of SA solutions to TSP (reproduced from ).
2.1.7 Central Force Optimization
Central Force Optimization (CFO)  is a new algorithm that departs significantly
from all other Nature inspired metaheuristics. ACO, PSO, SA, and the other algorithms
described below are all inherently stochastic. Every run with the same setup
parameters in general produces a different set of solutions. No two runs yield the same
results because these algorithms rely on true randomness in their functioning. The
values of certain key variables in the algorithm are, by definition, random variables that
are computed from a probability distribution. The values of these variables must vary
from one calculation to the next, and their values are completely unknown and
unknowable until the probabilistic calculation is performed.
CFO is quite different. It is based on an analogy drawn from gravitational
kinematics, which in turn is based on Newton’s laws of gravity and motion. Newton’s
laws are mathematically precise (completely deterministic) and, as a result, so too is
CFO. CFO searches the decision space by “flying” probes through it whose trajectories
are computed by deterministic equations that analogize Newton’s laws of motion.
Figure 5 shows how CFO’s probes move through a 3D decision space at each time step
sampling the decision space by computing the fitness of the function to be maximized
(shown by the darkened circles). CFO thus provides some major advantages over
stochastic algorithms, viz, every run with the same setup returns exactly the same
answers, and because of that characteristic only one run is necessary (stochastic
algorithms usually are run many times and the results averaged). CFO has been
effectively used for antenna optimization, and it holds considerable promise for use in
set‐top antenna design.
Figure 5. Central Force Optimization metaheuristic (reproduced from ).
2.1.8 Invasive Weed Optimization
Invasive Weed Optimization (IWO)  draws its inspiration from the colonization
characteristics of invasive flora as understood from weed biology and ecology. Like
ACO, PSO, GA, and SA, IWO is a population‐based stochastic algorithm. Weeds exhibit a
very strong tendency to opportunistically occupy (colonize) the interstitial spaces is a
cropping field. Spaces not occupied by crops, which usually do not spread, become
weed‐filled, and the weed then grows and propagates by consuming unutilized
resources in the field. The weed that uses these resources most effectively becomes the
dominant (fittest) weed. When a weed flowers, it produces seeds that are randomly
dispersed throughout the field until all interstitial space is occupied and all resources
Figure 6 shows a flow chart the IWO implementation used to solve
electromagnetic problems in . This flowchart starts out much the same as the GA
flowchart with a randomly generated population whose fitness is evaluated in the initial
step. Each weed then produces a number of seeds (reproduction) based on its fitness,
with weeds having better fitnesses being allowed to produce more seeds. The seeds are
then randomly dispersed through the decision space using a normal (Gaussian)
distribution of random numbers with mean value equal to the weed’s location. After
the new seeds have been dispersed, they are allowed to grow into new flowering
weeds, and the process is repeated until a convergent solution is generated. Because
the number of weeds grows constantly, a maximum weed population serves as a ceiling
on weed count. Whenever it is exceeded, the bottom worst plants are “weeded out” by
Figure 6. Invasive Weed Optimization flow chart (reproduced from ).
2.1.9 Intelligent Water Drop Algorithm
The Intelligent Water Drop algorithm (IWD) , like SA and CFO, analogizes a
physical process. But, like SA and unlike CFO, it is stochastic in nature instead of
deterministic. IWD is inspired by the notion that the seemingly random meanders in a
river or stream bed are, in fact, based on mechanisms that can be applied to effectively
solve optimization problems. Two principal factors are considered in IWD: water
velocity and soil characteristics. Each IWD flows from a source to a destination, initially
with non‐zero velocity and zero soil. Along its route, velocity and soil both may be
gained and lost. Soil inhibits drop velocity, so that between points the IWD’s velocity
increases inversely with soil (in a non‐linear manner). Figure 7 shows typical IWD results
for the Travelling Salesman Problem (see also discussion above on SA and TSP).
Figure 7. IWD results for the Traveling Salesman Problem (reproduced from ).
2.1.10 Bacteria Foraging Optimization
Bacteria Foraging Optimization (BFO)  mimics the natural behavior of bacterial
seeking food. The motion of individual bacteria is driven by avoiding noxious elements
in the environment while “swimming” upward along the food concentration gradient
hopefully locating a denser source of food. Chemotaxis is the process by which a
bacterium tumbles to orient itself, swims a fixed distance, and samples the food
concentration at its new location. If the concentration is greater than at the previous
location, then the bacterium continues in the same direction for another step. But if the
concentration is lower, then the bacterium tumbles into a new direction and explores it
instead. Each bacterium has a finite lifetime that limits the number of steps it can take.
At the end their lifetime bacteria that are in the highest food concentration regions are
allowed to “reproduce” by splitting into two new bacteria, while those in other regions
die. Because the bacteria are dispersed, after the reproduction step, only the best half
are retained, while the others die, thus preserving the total number of bacteria. The
user specifies the maximum number of chemotactic and reproductive steps, maximum
swim length, and maximum swims in a given direction.
2.1.11 An Antenna Optimization Example
This section presents a specific example of an optimized antenna design, a Self‐
structuring antenna (SSA) that can alter its electrical shape in near‐real time in response
to factors such as a changing environment or a change in mission. Typically these
structures comprise wire segments that can be interconnected using control signals
from a micro‐controller device. An example of a SSA that was optimized using three
different optimization algorithms appears in Figure 8 . The antenna in this case is an
asymmetric wire array containing 39 switches resulting in 549 billion possible antenna
The specific problem addressed in  was whether or not the optimization algorithms
would provide better performance than a simple random search for a configuration that
met specific performance criteria.
The wire structure was modeled with the Numerical Electromagnetics Code
(NEC) following all modeling guidelines with respect to wire segmentation and segment
length to diameter and wavelength ratios. The objective was to obtain low VSWR (< 2)
values at frequencies of 40, 100 and 400 MHz. The antenna grid measured 0.6 meter
square ( 08.0 on a side at 40 MHz), which electrically is quite small. Figures 8, 9 and 10
compare the results of a random search to those produced by the three optimization
algorithms. Random search performed very poorly at the lowest frequency, while each
of the optimization algorithms performed relatively well. The GA produced the
“tightest” results at 40 MHz (minimum standard deviation), and achieved the design
objective in 60% of the trials. This example shows that real‐time‐optimized SSAs are
within reach for the set‐top antenna application at this time, and that fairly economical
designs may be possible.
Figure 8. SSA geometry (reproduced from ).
Figure 9. SSA VSWR, random search [left], ACO [right] (reproduced from ).
Figure 10. SSA VSWR, SA [left], GA [right] (reproduced from ).
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1, nos.1/2, p. 71, 2009.
 Datta, T., and Misra, I. S., “Adaptive antenna Array Processing: The Bacteria‐
Foraging Algorithm and Particle‐Swarm Optimization,” IEEE Antennas and
Propagation Magazine, 51, no. 6, December 2009, p. 69.
 Coleman, C. M., Rothwell, E. J., and Ross, J. e., “Investigation of Simulated
Annealing, Ant‐Colony Optimization, and Genetic Algorithms for Self‐
Structuring Antennas, IEEE Trans. Antennas and Propagation, 52, 4, p. 1007,
2.2 Fragmented Antennas
As an additional illustration of the power of an electro‐magnetic (EM) optimization
algorithm, this section describes the computer program FAopt that begins with a grid of
wires as shown in Figures 1 and 2 and optimizes the placement of these wires using a
Binary Genetic Algorithm (BGA) and the NEC‐4  Method of Moments (MoM)
FORTRAN code. Each bit in the BGA chromosome is either a 1 or a 0, representing the
presence or absence of a wire, respectively. Quadrant 1 and 3 are a mirror image of
each other as well as quadrant 2 and 4, so that the antenna is symmetrical about its
diagonal. The resulting antenna is comprised of a series of plated conductors, some of
which are connected to one another. Some are capacitively/inductively coupled and act
as parasitic elements to increase the element impedance and radiation pattern
bandwidth. These types of radiating elements, when implemented as pixels rather than
wires, (see Maloney et al of GTRI ) are generally called “fragmented” antennas .
FAopt allows the user to choose the polarization as either vertical or horizontal, and the
antenna can be either directional or omni‐directional. The user also can choose the
dimensions for the desired antenna, as well as the frequency range. This program was
used to design a set‐top antenna to operate in the VHF and UHF bands. (See Section
3.0). During optimization a window is shown that updates the progress of the optimizer
with the best “figure of merit” (FoM) displayed along with the standing wave ratio.
Figure 1: FAopt Code’s GUI
Figure 2: Wire Grid
As an example, a thin planar directional antenna was optimized using FAopt as a
horizontally polarized antenna from 2400‐2480MHz with 8 blocks per row/column per
quadrant. The orientation appears in Figure 1, and the antenna model is shown in
Figure 3. The dimensions for this antenna are 2” by 2”. A prototype was built as shown
in Figure 4.
Figure 3. Fragmented Directional WiFi Antenna
Figure 4. Fragmented Directional WiFi Antenna
The simulation results, which were confirmed by measurements, shows the average
gain of this antenna to be better than 1.9 dBi. This antenna is a directional WiFi antenna
optimized to have a F/B ratio of about 6 dBi. Its radiation pattern is very consistent
across the band as shown in Figure 6, and its measured VSWR is better than 2:1 across
the band. Further work, such as changing the grid template or the optimization routine,
could be undertaken to make this approach more efficient for designing set‐top
Figure 5. Pattern of Directional WiFi Antenna
Figure 6. VSWR of Directional WiFi Antenna
The F/B ratio of simulated and measured are in fair agreement as shown in Figure 7.
Figure 7. F/B Ratio of Directional WiFi Antenna
Directional WiFi Antenna
VSWR of Directional WiFi Antenna
2400 2420 2440 2460 2480
F/B Ratio of Directional Wifi Antena
2400 2420 2440 2460 2480
 G. J. Burke, “Numerical Electromagnetics Code‐NEC‐4 Method of Moments”,
Lawrence Livermore National Laboratory, 1996
 J. G. Maloney, M. P. Kesler, P. H. Harms, G. S. Smith, United States Patent No. US
6323809 B1, Nov 27,2001
 B. Thors, H. Steyskal and H. Holter, “Broad‐Band Fragmented Aperture Phased
Array element Design Using Genetic Algorithms”, IEEE Trans. Antennas Propag.,
Vol 53, no. 10, pp. 3280‐3287
2.3 Non‐Foster Impedance Matching
Foster’s reactance theorem, which dates to 1924, states that a lossless reactance must have
a positive slope with frequency. Any lossless matching network presumably satisfies the
theorem, but a great deal of research that has been done on “non‐Foster matching” in the
past 15 years suggests otherwise. This section examines how non‐Foster networks may be
useful in set‐top antennas to match an electrically small element (characterized by a low
real and high negative imaginary complex impedance) within, for example, the low‐VHF
Until recently no one has created a practical antenna system using this method that showed
much of an improvement in performance. This was due to limitations in the semiconductor
technology at the time that permitted only narrow band solutions. Semiconductor
technology has improved considerably in recent years resulting in increased bandwidth,
lowered noise, and decreased losses in the devices. Electrically small antennas present high
Q impedances characterized by large reactances and small radiation resistances. For such
antennas, the effectiveness of passive matching circuits is severely limited by gain
bandwidth theory, which predicts narrow bandwidths and/or poor gain. The result
inevitably is a poor signal to noise ratio compared with a larger antenna.
Non‐Foster matching uses negative impedance converters (NIC) to create “negative”
capacitors and “negative” inductors [1‐2]. It is possible, for example, to use negative
capacitance to cancel out the reactance in a short dipole or monopole. This leaves only a
small real impedance which could then be matched to, say, 75 Ω using a transformer. One
disadvantage of this approach is that transformers add to the antenna size which could be
undesirable. An alternative to matching could be achieved by placing the non‐Foster
matching circuit away from the feed to control both the real impedance as well as the
reactance of the antenna . The potential improvement in antenna performance is very
significant. Theoretical bandwidth improvements have been shown for a small loop
(increasing bandwidth from 50 MHz to 300MHz at a center frequency of 700MHz), as well
as a small broadband dipole (increasing bandwidth from 250MHz to 1000MHz, lowering the
lower operating frequency from 250 MHz to 190 MHz).
It should be noted that the Figures and Tables in this section were taken from references
2 through 5, listed at the end of this section.
Figure 1. Linvill’s ideal OCS NICs terminated in a resistance.
Fiugre 2. Linvill’s ideal SCS NICs terminated in a resistance.
With an ideal transistor, a pure negative resistance is achievable. In Linvill’s actual
realizations (Figures 1 and 2), a substantial reactive component of the input impedance Z
accompanies the negative resistance, resulting in a low Q, where the “resistive” Q is defined
as: Q=Re(Z)/Im(Z). The circuits shown are “open‐circuit stable” (OCS), meaning, practically,
that if a very large resistance terminates the negative‐resistance one‐ports on the left, then
the overall network will be stable. The networks of Figure 1 can be turned around as shown
in Figure 2, where negative resistances of the “short‐circuit stable” (SCS) variety are
obtained. Again, practically speaking, this means these networks will be stable if a very
large conductance is placed across the input. An initial step in improving the power
efficiency of non‐Foster matching is simply to use a single‐transistor version of the NIC .
Figure 3. Non Foster Matching
An example of a non‐Foster matching circuit is shown in Figure 3. Although not shown in
the schematic, a transformer would be used at the input to make the effective value of
Rsource, as seen by the matching circuit, to be 18.75Ω (4:1 impedance ratio).The circuit
negates antenna’s Capacitance Cant and the voltage‐managing resonating inductor Lres, as
well as the radiation resistance r. The idea is to cancel the negated Lres and Cant with C and L
on the left, respectively. The negated r is absorbed by the much larger source resistance
and does not affect operation of the circuit. The source resistance would be 75 ohms in this
An example of non‐Foster matching for a 12 inch dipole is shown in Figure 4. It is shown in
Figure 5 that this non‐Foster matching network creates a negative capacitance that is used
to cancel out the reactance of the 12” dipole. This is used along with a transformer to help
match the radiation resistance of the antenna to the source impedance of 50 Ω. The non‐
Foster matching network is compared to a similar dipole with a lossy passive network for
improvements in signal to noise as shown in Figures 6 and 7. The improvement in SNR is
quite remarkable, as much as 25 dB around 44 MHz.
Figure 4. Non‐Foster Matching Network for a 12” Dipole.
Figure 5. Measured and simulated results for capacitance of non‐Foster matching network.
Figure 6. Setup for Signal to Noise improvement measurement.
Figure 7. Measured Improvement of Signal to Noise
Non‐Foster matching may be particularly useful for the set‐top application because it works
best with electrically small antennas. Therefore, this type of match could be useful when
designing a small VHF/UHF DTV set‐top antenna. Improvements in signal to noise
measurements have been shown from 20MHz to 400 MHz . In particular, the use of non‐
Foster techniques to impedance match a lossy electrically‐small dipole antenna has been
quite effective. On an antenna range, there was a measured 30 dB gain improvement over
60 – 200 MHz, with several dB of improvement as high as 400 MHz. Again, the comparison
was to the same antenna with no matching at all. Although no S/N measurements were
made, the circuits used were based upon the same low‐noise designs developed earlier for
lower‐frequency circuits. Because of the lossiness of the antenna, passive matching can do
a little better than no matching at all, and these results are illustrated in simulation. This
computer simulation designed a number of “best‐effort” passive matching networks and
calculated the transducer gain (S21) between a 50 ohm source and the complex impedance
of a 6‐inch monopole. S21 for no matching and S21 for a single, ideal negative capacitance
whose value (‐5.54pF) exactly cancels the antenna reactance at 30 MHz, were calculated.
Plots of the real and imaginary parts of the antenna impedance are shown in Figure 8. The
various matching networks are illustrated in Figure 9; each of these would appear in place
of the NIC in Figure 10b.
Figure 9‐A. Real Part of Antenna Impedance for a six inch monopole
Figure 9‐B. Imaginary Part of Antenna Impedance for a six inch monopole
Figure 10. Various Matching Techniques for six inch monopole
Table 1. Computed Average Gain of all matching techniques.
Figure 11. Plots of Computed Gains of all matching techniques for a six inch monopole
The six inch monopole was then built using both the non‐Foster matching network and an
unmatched antenna. They were both measured with the results shown in Figure 12. In the
design, NE68019 BJTs were used in the NICs because these devices provided good overall
performance over the entire frequency band. Again, the results are quite remarkable, with
the non‐Foster antenna showing average gain improvements between about 2 and 28 dB.
It is noteworthy that the greatest improvements are at lower frequencies, between about
60 and 210 MHz, approximately the low and high VHF bands.
Figure 12. Gain of a non‐Foster matching network over the same antenna with no
Figure 13. Plots of Computed Gains of all matching techniques for a six inch dipole
Results for an electrically small lossy dipole are shown in Figure 13 (this antenna was
simulated, not measured). The actual gain may be less due to noise of the device if this
were to be built and tested. Thus, realization of non‐Foster technology is still limited by the
analog circuitry performance. With respect to the set‐top application, it is very reasonable
to expect that as better silicon devices are developed covering the entire television
frequency range will be possible with substantially better antenna performance in terms of
gain and signal‐to‐noise ratio.
 R.C Hansen, “Electrically Small Super directive and Super conducting Antennas”, John
Wiley & Sons Inc. 2006
 J. T Aberle, R. Leopsinger‐Romak, C. A Balanis, “Antennas with Non‐Foster Matching
Networks”, Morgan & Claypool Publishers, 2007
 S. Koulouridis and J.L. Volakis, “Non‐Foster circuits for small broadband antennas”,
 S. E. Sussman‐Fort and R. M. Rudish “Non‐Foster Impedance Matching of Electrically‐
Small Antennas”,IEEE Transactions on Antennas and Prop., Vol 57,NO. 8, August 2009
 S. E. Sussman‐Fort and R. M. Rudish, “Non‐Foster impedance matching of a lossy,
electrically‐small antenna over an extended frequency range” presented at the Antenna
Applicat. Symp., Allerton Park, IL, Sep. 2007
2.4 Active RF Noise Cancelling
Considerable work has been done to mitigate noise from devices near receiving antennas,
as is particularly useful in set‐top antennas. An antenna placed on top of a television is
especially vulnerable because it may pick up noise from internal circuitry. But this noise can
be reduced, which in turn reduces the noise floor, or, equivalently, increases the signal‐to‐
noise ratio (SNR). Intersil  has created the QHx220 chip that accomplishes this e.g. at
UHF frequencies, or more precisely from 300 MHz up to >3GHz. MegaWave has evaluated
the Intersil noise cancelation chip QHx220 shown in Figure 1. The chip was tested for signal
to noise improvements at 535MHz as shown in Figure 2. Measured results showed an
improvement of about 12dB in SNR as shown in Figures 3 and 4. This technology has been
extended down to FM covering the VHF III band , which is necessary for this approach to
be viable in the set‐top environment.
Figure 1: Intersil’s Noise Cancellation Chip QHx220
This system works on a principle similar to Bose and Sony noise cancelling headsets, but at a
~1,000,000 times higher frequency. It requires an I and Q setting obtained from the DTV
receiver’s channel setting and the system’s link quality parameter (BER, SNR, RSSI, etc.). This
is done on either a micro‐processor or in the baseband processor, which run a set of small
algorithms. As a result the noise minimum is centered in the desired TV channel and signals
‐ formerly buried in the noise floor ‐ are being restored (Figure 4, e.g. at 549MHz).
Figure 2 Noise Cancellation Test and Evaluation Setup
Figures 3‐4: Measured results before and after noise cancelation
 Intersil Corporation, 1001 Murphy Ranch Rd, Milpitas, Ca 95035, www.Intersil.com
 QHx120 (development) works from 75MHz up to 225MHz.
2.5 Automatic Antenna Matching Systems
From an impedance matching point of view, the ideal antenna has a perfectly
flat, purely resistive input impedance across its entire operating frequency range. The
value of that impedance should be 70 Ω because this is the nominal industry standard
characteristic impedances of the coaxial cable used in television receivers. Of course, no
antenna is perfect. Quite to the contrary, most antennas’ impedance variations with
frequency are usually quite dramatic. This is particularly true for electrically small
antennas, which tend to exhibit low radiation resistances and very high reactances in a
narrow frequency range. Set‐top DTV television antennas are not necessarily electrically
small in the high VHF and UHF bands, but are at low VHF. Due to their required
bandwidth (54‐698 MHz) they invariably exhibit wide impedance fluctuations.
An impedance mismatch leads to losses and reflected energy that is not
transferred to the receiver. These inefficiencies can be mitigated to a great degree by
precise impedance matching that is provided by an “antenna tuner” (AT) connected
between the antenna output terminals and the receiver input port. The AT may
comprise a network of discrete components, or it may be a distributed, continuously
varying structure (a tapered transmission line, for example), or some combination of
Lumped element matching networks go back to the earliest days of radio, and
they often require manual adjustment of the matching elements (usually variable
capacitors or roller inductors). In the early 1960s the military developed “ALE”
(automatic link establishment) systems that employed remotely controlled automatic
antenna tuners. These networks are usually located at the antenna feed and remotely
tuned automatically based on VSWR (voltage standing wave ratio) measurements made
at the transmitter. Essentially the same approach is taken with the built‐in automatic
ATs that are common in modern transmitters (most amateur radios contain built‐in auto
ATs, for example).
Modern low‐power ATs use relays to switch lumped elements (inductors,
capacitors) in and out of the matching network until the desired match is achieved. This
technology is highly developed and readily available for use in set‐top TV receive
antennas. It therefore is not reviewed in this report. Instead, the emphasis here is on
new, developing technologies that may be useful for antenna impedance matching.
Four broad categories have been identified.
(a) Voltage controlled reactive elements are capacitors and inductors whose
values are controlled by an applied voltage. Instead of switching discrete components
in and out of a matching network using relays, the same operation is accomplished by
varying the voltage applied to the reactive element. This type of non‐mechanical
system provides faster response time and more continuous control in a less noisy and
probably smaller space than a relay‐based AT.
(b) Integrated ATs are “antenna tuners on a chip,” a complete device that is
fabricated on a single chip or in a very small module with only a few external
components. Integrated ATs have received considerable attention for cellular
applications, and they work quite well. Chip‐level devices successfully match up to 10:1
VSWRs in the highly fluctuating cellular antenna environment with rapid response times
and high efficiency. These devices are constantly improving, and it is likely that existing
technology can meet set‐top AT requirements.
(c) Tunable tracking filters are already used across the VHF/UHF TV bands
for wideband DTV (digital television) tuners, and they work quite well, providing very
flat passbands, steep skirts, and harmonic suppression from the tuner’s local oscillator.
Extremely small devices have been designed and computer‐simulated, and some
measured data are available confirming achievement of design objectives. This is an
evolving technology that may be transferrable to set‐top AT requirements.
(d) Software defined radio (SDR) is a new concept in wireless
communications that seeks to move system properties from the hardware layer to a
high‐level software layer by developing hardware modules that are fully software‐
controllable and configurable. SDR radios are gaining wide acceptance in many
applications, and it is a developing technology. SDR “smart antenna” technology, which
may be applicable to the set‐top environment, consequently is a candidate technology
for a television AT.
126.96.36.199 Voltage Controlled Reactive Elements
(a) BZN (pryrochlore bismuth zinc niobate) is a non‐ferroelectric dielectric
material that shows great promise for a new class of voltage tunable thin film capacitors
. When it was introduced in 2006, BZN exhibited the lowest loss of any room
temperature complex oxide film with a very high dielectric constant and high tunability
of (loss tangent ~0.0005, dielectric constant ~180, ~55% tuning range @ 1 MHz). Losses
increase in the microwave range, but are controllable by modifying the electrode
Figure 1. BZN capacitor high‐frequency measurements (reproduced from )
Measured data for different size BZN thin film capacitors at higher frequencies
appear in Figure 1. For small devices (100 μm2
) the Q‐factor exceeds 200 up to 20 GHz
(de‐embedded data, right plot). Through about 3 GHz Q ~ 1000. In addition, self‐
resonance in the BZN thin film structure is well above 20 GHz. By comparison, the best
state‐of‐the‐art thin‐film BST (barium strontium titanate) devices in 2006 exhibited Qs
that decreased monotonically from about 100 at 1 GHz to around 20 at 20 GHz. BZN is a
far superior tunable dielectric. BZN and similar materials (see below) hold considerable
promise for the development of completely solid‐state ATs that can be used to tune set‐
top TV broadcast band antennas.
(b) PZN (lead zinc niobate) thin film interdigital capacitors (IDC) have been
fabricated to increase tunability and reduce bias voltage compared to BZN IDC
implementations. Cubic pyrochlore PZN thin film dielectrics provide superior
performance through the microwave range. Figure 2 shows a typical configuration. The
IDC is fabricated on a silicon substrate in the usual coplanar waveguide (CPW)
configuration. Figure 3 plots the measured low‐frequency dielectric constant (blue)
which increases from just over 200 to about 230 at 10 MHz. The tenability as a function
of applied bias voltage decreases from unity to about 0.75 with an applied electric field
of 2 mV/cm (bias voltage of 5.5 volts). At 1 GHz the device Q falls to about 10 at all bias
voltages, but the tunability remains at 26% (compared to 25% at 1 MHz) for 5.5 volts
applied bias. The PZN tunable capacitor thus is a strong candidate for voltage‐tunable
set‐top television antenna ATs, and its performance is comparable or better than that of
Figure 2. PZN IDC structure (reproduced from ). Figure 3. PZN IDC performance
(reproduced from ).
(c) DuNE™ capacitors are a new patent‐pending technology developed by Peregrine
Semiconductor  for DTCs (Digitally Tunable Capacitors). The manufacturing process is
based on Peregrine’s proprietary UltraCMOS™ technology supplemented by its HaRP™
design methodology. UltraCMOS™ is a patented low‐power SOI (silicon on insulator)
variant chip architecture that reduces transistor capacitance resulting in increased
switching speed. HaRP™ is a design technology that produces significant improvements
in harmonic results, linearity and overall RF performance.
The company claims that its DTCs provide digitally controllable capacitances in
the range 0.5 to 10 pF with either 3:1 or 6:1 tuning ratios and 32 addressable states (5
bits). Qs range from 40 to 80 between 1 and 2 GHz with device switching times below 5
μsec. Power handling capability is quite good (+38 dBm @ 50 Ω), while power
consumption is low (100 μamps @ 2.4‐3.5 volts DC). The manufacturer claims its DTCs
provide better performance than currently available MEMS (microelectromechanical
system) or BST devices, and they are readily available as off‐the‐shelf components.
The digitally tuned capacitor schematic diagram and its “flip‐chip” package
appear in Figure 4. The DTC acts as a series capacitor with two RF terminals. It is
powered by a single DC line, and controlled by a three‐wire serial interface. The control
word is five bits long. The device is extremely small (1.26 mm x 0.81 mm) and thus well‐
suited for use in TV set‐top antennas. Measured performance is shown in Figure 5.
Capacitance variation is very linear with control state (32 increments), and the variation
ratio is 3:1, from 1.15 pF to 3.4 pF. The DTC Q‐factor at 900 MHz, a representative
cellular frequency, varies from about 53 to about 72, which is quite good.
Figure 4. Digitally Tuned Capacitor (reproduced from ).
Figure 5. DTC measured performance (reproduced from ).
DTCs have been effectively applied in cellular phone ATs. A typical configuration
is shown in Figure 6. The transceiver, in this case a cellular handset, is connect to its
antenna through an AT comprising four main elements: (1) serial interface; (2) digital
mismatch sensor; (3) tuning algorithm; and (4) DTC core. This structure is
representative of ATs based on other tuning technologies besides DTCs. For example,
this same architecture would be used with MEMS‐based or tunable thin film capacitor
ATs. Note that the diagram does not show the RF path between the antenna and
transceiver, only the AT control data path. The serial interface exchanges digital data
with the transceiver on the AT status. The mismatch sensor responds to reflected
power resulting from any impedance mismatch between the antenna and the system’s
characteristic impedance. Digital mismatch level data are processed by the tuning
algorithm that determines which DTCs should be activated and at what capacitance
level. Control words are sent to the DTCs in the matching network (DTC core) to effect
the impedance match between the antenna and transceiver.
Figure 6. DTC‐based AT (reproduced from ).
(d) Dual‐gap Tunable MEMS Capacitors are made using a novel fabrication process
that creates two gap device that has an extremely wide tuning range (as much as a
factor of 15) with high Q . Figure 1 shows the dual‐gap structure schematically and as
implemented using a “two hump” sacrificial layer which, when removed, creates two
MEMS gaps instead of one. In the schematic, the two moveable electrodes are
indicated by the double downward arrows. As these suspended elements move closer
to the substrate electrodes the capacitance increases. The structure is a MEMS device
designed for a linear capacitance variation with applied control voltage.
In the fabricated prototype, the central region (Ec) is 250μm x 80μm (LxW) with a
1.5μm fixed gap. The two “beams” are 800μm long by 80μm wide creating a variable
gap because they are movable. The “actuation area” patches (Ea) are 200μm long and
80 um wide with a gap of 4.5μm with the beams not deflected. This structure produces
a minimum capacitance of 0.12 pF that is voltage controllable to a maximum
capacitance of 1.77 pF. The resulting tuning range is %1375
, which is quite
remarkable. The required bias (“pull in”) voltage is less than 12 volts. This device is not
commercially available, but it represents a class of tunable devices that should be very
useful in the set‐top AT when (and if) they become COTS (commercial off‐the‐shelf)
Figure 7. Dual‐gap tunable MEMS capacitor (reproduced from ).
(e) MEMS floating dielectric capacitors (MFDCs) are a recent promising
development . The movable dielectric is a new actuation principle in which a floating
movable dielectric is electrostatically maneuvered to vary the capacitance. Figure 8
shows the new concept schematically. A mechanical spring returns the movable
dielectric to its undisturbed position when no force is applied by the electrostatic comb
drive. When the dive is activated, the dielectric moves closer to the top and bottom
capacitor plates thereby increasing the capacitance. Preferentially the RF path is
through the plates, not through the dielectric and spring, which increases losses.
Figure 8. MFDC concept (reproduced from ).
Figure 9 shows schematic RF signal path superimposed on the MFDC’s actual
structure as fabricated. In the left pane the signal flows through the spring, which
increases the capacitance at the expense of increased losses. In the right pane the
signal flows between the top and bottom plates through the movable dielectric. This
configuration results in lower losses, but lower capacitance as well. Of course, either
configuration can be used, depending upon the specific application.
Figure 9. MFDC as fabricated and measured data (reproduced from ).
Figure 10. MFDC as fabricated and measured data (reproduced from ).
Figure 10 provides a scanning electron (SEM) microscope image of a fabricated
MFDC (the area outlined in orange, lower pane) and measured performance data.
Comparing Figs. 9 and 10, the features in the schematic are readily identifiable in the
SEM photo. The actuator pads are labeled D+ and D‐ in the photo, while the spring is
attached to pad C. Pads A and B are the connections for their associated comb plates.
The applied bias voltage ranged from ‐120 VDC to +120VDC, resulting in capacitance
ranges of about 760 fF (femtoFarad) to 2100 fF for RF signals passing through the spring
element (pink curve). The initial capacitance (no bias voltage) was 830 fF, leading to a tuning
range of approximately 170%. The 1 GHz Q‐factor was 0.35, which is low by comparison to
other technologies. When the RF path is through the movable dielectric (blue curve),
capacitance ranged from just over 100 fF to about 550 fF yielding a tuning range of
approximately 370% with an initial capacitance of 135 fF. As expected, the reduced loss
resulting from routing the signal away from the spring results is a much better Q‐factor of 56 at
1 GHz. Overall, this performance is not as good as that provided by other technologies, but
MFDCs are a new concept that requires further development. The MFDC approach certainly
merits consideration as a potentially useful future technology for the set‐top AT.
(f) The voltage controlled semiconductor inductor (VCSI) is another reactive
component that should be useful in set‐top ATs. Inductor values are usually fixed, so
that obtaining specific value of inductance in an AT is usually accomplished by
mechanically switching small inductors in and out of the matching circuit using relays or
MEMS RF switches. Recently disclosed VCSI devices  should be particularly useful at
broadcast TV frequencies. The device addresses the problem of limited tuning range
provided by voltage tunable capacitors by varying the inductance instead.
Generally is comprises regular coil turns of wire interconnected by
semiconductor diodes that can connect individual turn to create a VCSI. Including a
resistor and capacitor creates a complete tunable RLC circuit. Figure 11 illustrates this
patented technology. The left pane shows a perspective view of the device, which
includes conductive loops (205) connected to a semiconductor bar comprising P‐ and N‐
type regions at each end (208 and 210, respectively) connected by a depletion region
(212). Applying a voltage across terminals LA and LB varies the length of the depletion
region which acts as an insulator. Because its size is proportional to the applied bias
voltage, individual coil turns are either connected or disconnected in proportion to the
voltage, thereby creating a voltage‐variable inductor.
Figure 11. Voltage controlled semiconductor inductor (reproduced from ).
A helical‐turn implementation of the VCSI is shown in the right pane of Figure 11.
The same semiconductor diode structure comprising P‐ and N‐type end regions
connected to a central depletion layer (DL) whose length is voltage‐variable. The figure
provides a schematic representation of how the three regions’ lengths vary with bias
voltage compared to the “off” state shown in the left pane. In the diagram, coil turns C
through F are electrically short‐circuited because they are outside the insulating DL. In
this case, only turns A and B are active elements in the inductor.
The VCSI may become an important element in set‐top ATs because it provides a
complete RLC tuning in a single chip‐level device. A bank of VCSIs, for example, could
comprise the switchable reactive elements in a set‐top AT that are completely voltage‐
controllable, thereby eliminating the need for mechanical switching relays. VCSI
therefore is an attractive emerging technology that merits watching.
188.8.131.52 Fully Integrated ATs
In addition to the emerging component‐level technologies described in §1, fully
integrated ATs have been developed that also merit consideration for set‐top tuners.
This section examines developments in that area.
(a) A reconfigurable RF‐MEMS‐based matching network is described in . The
chip‐level device’s circuit diagram is shown in Figure 12. It comprises two stages, the
first of which is a Pi‐match section with four shunt capacitor‐series inductor (CL)
sections. A total of eight RF‐MEMS switches are employed yielding 28
shunt capacitors are formed from bi‐valued MEMS varactors. The variable capacitor in
series with each fixed inductor, also a MEMS varactor, has the effect of adjusting the
series inductance. The second section is a phase‐shifter comprising a 3‐dB 90‐degree
coupler connected to a reflective load. MEMS varactors adjust the load reflection
coefficient to control the overall phase shift. A total of 23
phase rotations are possible in
the phase shifter, resulting in as much as 340 degrees of total phase shift that can be
applied to the impedances at the output of the Pi‐match. The network is designed to
work at a 50 Ω impedance level.
Figure 12. Reconfigurable RF‐MEMS‐based matching network circuit (reproduced from
The chip layout and fabricated device are shown in Figure 13 with the major
sections being labeled on the layout diagram. The chip area is extremely small, only 40
(slightly larger than 3x12 mm). The published report indicates that the chip has
been fabricated and was undergoing testing, but no actual measured data were
reported. Instead simulated performance at a single frequency (620 MHz) was
calculated at each of the possible 2,048 impedance combinations and plotted on a
Smith chart as shown in Figure 14. The circles are impedance values computed for the
first stage alone, while the dots represent the show the performance of the complete
reconfigurable matching circuit. The fairly uniform distribution of dots throughout the
Smith chart suggest that the matching circuit will effectively match an extremely wide
range of impedances to 50 Ω. This technology is directly applicable to the set‐top AT
and appears to be on the verge of realization. Future published results for
reconfigurable RF‐MEMS‐based matching networks clearly bear watching.
Figure 13. Reconfigurable RF‐MEMS‐based matching network layout & fabrication
(reproduced from ).
Figure 14. Reconfigurable RF‐MEMS‐based matching network performance (reproduced
(b) The AT using only RF signal amplitudes described in  may be useful for the set‐
top application because it specifically addresses the issue of highly variable,
unpredictable and uncontrollable environments. This new technology was developed
for mobile applications such as cellular transceivers because their fluctuating
environment often causes VSWR spikes approaching 10:1 in nominal 50 Ω systems.
These fluctuations are frequently transient on time scales in the millisecond range. The
new AT design accommodates this environment, and it may be useful in the highly
variable TV set‐top antenna environment as well.
The architecture appears in Figure 15, which includes schematic Smith chart
representations of the impedance at various points. ATs generally comprise an
impedance sensor, a tunable matching network, and control circuitry that changes the
network parameters in order to achieve an acceptable VSWR (usually relative to 50 Ω
with VSWR < 2‐3:1). The general approach shown in Figure 15 therefore is applicable to
any matching network. The new matching concept involves a two‐step process: (1) the
reactance is essentially tuned out using a series or shunt reactance; and (2) a tunable
“transformer” changes the remaining resistive component to the desired value of
Figure 15. New AT architecture for fluctuating environment (reproduced from ).
In the new AT, the antenna impedance sensor makes a quasi‐DC log‐peak
measurement of the RF signal amplitude at three points in the signal chain (V1, V2, V3 in
Figure 16). Details of the log‐peak detector circuit are shown in Figure 17. These data
combined with the known transfer functions of the reactive elements (jXext) permit a
calculation of the impedance’s imaginary part. The real part of the impedance is
determined from the signal level VΔR and an offset voltage that brings this level to a
target value corresponding to 1:1 VSWR. The tunable transformer comprises a T‐
network lumped transmission line made of two adjustable inductors and an adjustable
capacitor. Any appropriate device can be used to implement these elements.
Figure 16. AT functional block diagram (reproduced from ).
Two demonstration versions of this new AT design was built and tested at
900 MHz. Measurements confirmed that they could reliably and quickly automatically
tune antennas with ]1588.155005[ jZin (10:1 VSWR). Although the sizes
were not reported, these demonstration units were intended for use in cellular
handsets, so that the size is certainly consistent with use in a set‐top television AT.
Figure 17. Details of log‐peak detector circuit in Figure 16 (reproduced from ).
184.108.40.206 On‐Chip Tracking Filters
While tracking filters are not AT’s, their technology likely is applicable to antenna
tuners ATs, and consequently should be monitored for application in set‐top devices.
This section describes three different types of tracking filter.
A device developed specifically for DTV tuners is described in . A new
architecture is proposed that includes two separate filters, a harmonic rejection tracking
filter (HRTF) and an RF tracking filter (RFTF) that are controlled by complementary digital
switches. The new architecture is shown in Figure 18. The entire device is fabricated as
a single integrated circuit (IC) chip. The filter stage is shown in blue located between the
low noise amplifier (LNA) connected to the antenna and the DTV tuner.
Unlike narrowband RF receivers, wideband DTV tuners are prone to interference
from local oscillator odd harmonics mixing with signals at the lower television channels
(48‐287 MHz). This problem is addressed by the HRTF that provides a high‐order band‐
pass tracking filter with 3rd
‐order harmonic rejection greater than 60 dB. On the UHF
television channels the RFTF tunes 287‐860 MHz with narrow band response (20 MHz at
‐3 dB). This stage is implemented using a cascade of tunable 2nd
Computer simulation of the new architecture’s performance predict 48‐860 MHz
operation with tunable adjustable bandwidth of 8‐20 MHz, 5‐15 dB N+2 channel
rejection (16 MHz offset), and 3rd
‐order harmonic rejection of 60 dB from the HRTF. The
RFTF stage provides 4.2 dB of N+2 channel rejection above 287 MHz. This device can be
fabricated on‐chip using 0.13 μm CMOS technology with 1.2 VDC supply voltage and
19.8 mA current draw (total of 24 mW power consumption).
Figure 18. New DTV tracking filter architecture (reproduced from ).
Another potentially important emerging filter technology is the use of low‐
temperature co‐fired ceramic (LTCC) technology to fabricate fully integrated multi‐layer
tunable filters for RF and microwave use. A typical device is shown in Figure 18. Passive
Figure 18. LTCC filter structure (reproduced from ).
Elements (resistors, inductors, capacitors) are integrated on the surface or embedded in
a multilayer substrate. Various configurations have been demonstrated, including
bandpass filters (380 MHz‐2.4 GHz), 3‐stage Butterworth bandpass filters (1.2 GHz), and
an electronically tunable microstrip combline filter. These prototype devices point to
LTCC technology’s utility for RF and microwave applications. At this point the new
multilayer architecture proposed in  that involves switching between layers for
tuning is being computer‐modeled, but working devices based on that approach have
not been fabricated. LTCC technology may be very attractive in the television frequency
range because of its potential for very high levels of integration resulting from the
A third example of an on‐chip tracking filter is provided by . A complete
tunable structure was fabricated and tested. This chip occupied an area of only 2.8 mm2
(fabricated with 0.18 μm CMOS) requiring 34‐120 mA at 1.8 VDC. A photomicrograph of
the chip appears in Figure 19, and its architecture in Figure 20. The device comprises
cascaded RLC sections as shown in Figure 20, each containing a digitally programmable
on‐chip capacitor and resistor, and an off‐chip fixed inductor. The resistor is adjusted
with 8‐bit resolution, while the capacitor uses a 10‐bit control signal.
Figure 19. Tunable LC‐Tracking filter as fabricated (reproduced from ).
Figure 20. Tunable LC‐Tracking filter architecture (reproduced from ).
Figure 21 shows the tracking filter’s measured performance data from 125 MHz
to 1.06 GHz. Its response in the 5.6 MHz passband is very flat with a ripple less than 0.2
dB. The noise figure in this device appears to be somewhat high (16.8‐19.5 dB), but the
third‐order intercept points are good (~128/167 dBμV, in/out of band). Frequency
selectivity is good at > 36 dB, and the power consumption quite low (<~0.2 W
maximum). This example shows that very effective single‐chip RF tracking filters with
minimal off‐chip components (in this case two inductors) can be designed and
fabricated for set‐top use using currently available technologies.
Figure 21. Measured performance of tunable LC‐Tracking filter (reproduced from ).
220.127.116.11 Software Defined Radios
A software defined radio (SDR) is an element of a wireless communication
network whose operational modes and parameters can be changed or augmented post‐
manufacture via software. The essential idea is that a flexible hardware layer exists
whose function can be controlled and modified entirely by a computer program, as
opposed to requiring hardware modifications of any kind. The SDR concept spans many
radio network technologies including cellular systems, personal communications
services (PCS), 3rd
generation wireless (3G and 4G), mobile data, emergency
services, paging, messaging, and military/government communications, and any future
modifications to these existing services or entirely new ones. The FCC (Federal
Communications Commission) definition is more restrictive in that it applies only to the
transmitter side of an SDR. But, as a practical matter, the SDR concept applies to any
wireless device whose characteristics are software‐controllable, whether it be the
transmitter, receiver, both, or some other element such as a modem.
SDR technology is relevant to the television set‐top AT because there is a
developing standard that specifically addresses the issue of “smart antennas” (SAs) in
the context of SDR. This type of antenna and its associated AT may be useful for the set‐
top application and consequently should be monitored as an emerging technology. The
high‐level SDR smart antenna architecture appears in Figure 22. The hardware layer
comprises M transmit antennas and N receive antennas because SDR in general
supports two‐way communication (in the step‐top TV receive application, of course,
there are no transmit antennas). Each antenna is has a separate RF/IF processing chain
with the smart antenna signal processing (“waveform application”) being applied to the
baseband signal. Device drivers in the middleware layer control various programmable
hardware devices, such as ASSPs (application‐specific standard processors), FPGAs
(field‐programmable gate arrays), DSPs (digital signal processors), and GPPs (general
Figure 22. SDR smart antenna high‐level architecture (reproduced from ).
Figure 23 illustrates a typical deployment of an SA API (application program
interface). The SA control device (for example, an antenna tuner, tracking filter, MEMS‐
based controller, fluidic element controller, and so on) is operated by a GPP controlled
by a CORBA interface (common object request broker architecture). The high‐level SA
algorithm controls appropriate CORBA‐compatible drivers for DSP baseband processing,
antenna control, and other functions such as synchronization and other SDR devices
that may be controlled by the SA algorithm. The basic concept in this structure is that
any hardware module involved in controlling the SA or processing its signal is
controllable by the API with variable parameters.
Figure 23. SDR Smart Antenna API deployment (reproduced from ).